Optical driver with active boost

ABSTRACT

The present disclosure provide for active boost in an electrical driver via a frequency comparator, configured to determine operational characteristics of an electrical circuit connected to an optical modulator based on a frequency difference between a ring oscillator and the clock signal; an electrical driver configured to drive a phase shift of a first optical signal carried on a first arm relative to a second optical signal carried on a second arm of an optical modulator, the electrical driver comprising: a first signal pathway, connected to the first arm of the optical modulator, wherein the first signal pathway includes: an adjustable gain inverter, electrically connected to first and second nodes; a fixed gain inverter, electrically connected to the first and second nodes; an inductor electrically connected between the second node and a third node; and a non-inverting amplifier connected between the third node and the first node.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of co-pending U.S. patent application Ser. No. 16/746,843 filed Jan. 18, 2020. The aforementioned related patent application is herein incorporated by reference in its entirety.

TECHNICAL FIELD

Embodiments presented in this disclosure generally relate to drivers for use with Mach-Zehnder Interferometers (MZI) and other optical modulators. More specifically, embodiments disclosed herein provide for the tuning of the driver to operate with lower power consumption while providing high bandwidth when desired via an active boost.

BACKGROUND

As the data rates carried in optical devices increase, the power needed to drive those devices increases. This additional power flowing in the optical devices can have significant thermal effects on the systems using such optical devices and may thermally degrade portions of the optical device itself—leading to a shorter lifetime of the device. To reduce the thermal effects, operators have reduced the amount of power used to drive the optical devices, which can lead to insufficient bandwidth to carry the desired signals.

BRIEF DESCRIPTION OF THE DRAWINGS

So that the manner in which the above-recited features of the present disclosure can be understood in detail, a more particular description of the disclosure, briefly summarized above, may be had by reference to embodiments, some of which are illustrated in the appended drawings. It is to be noted, however, that the appended drawings illustrate typical embodiments and are therefore not to be considered limiting; other equally effective embodiments are contemplated.

FIG. 1 is a block diagram of a photonic platform, according to embodiments of the present disclosure.

FIG. 2A illustrates a multi-segment electro-optic modulation device, such as may be used as a modulator, according to embodiments described herein.

FIG. 2B is a schematic view of a modulator segment coupled with outputs of an exemplary modulator driver, according to embodiments described herein.

FIGS. 3A and 3B are block diagrams of a driver, according to embodiments described herein.

FIG. 4 is a block diagram of an exemplary conversion stage of a modulator driver, according to embodiments described herein.

FIGS. 5A and 5B provide a schematic view of an input stage of a driver, according to embodiments of the present disclosure.

FIG. 6 illustrates a schematic view of a buildup stage including a chain of buffers arranged in series across a first arm and a second arm, according to embodiments of the present disclosure.

FIG. 7 illustrates a schematic view of an individual non-inverting feedback buffer, according to embodiments of the present disclosure.

FIG. 8 illustrates a schematic view of an individual inverting feedback buffer, according to embodiments of the present disclosure.

FIGS. 9A and 9B illustrate eye diagrams, according to embodiments of the present disclosure.

FIG. 10 is a flowchart of a method for tuning a driver for a modulator, according to embodiments of the present disclosure.

To facilitate understanding, identical reference numerals have been used, where possible, to designate identical elements that are common to the figures. It is contemplated that elements disclosed in one embodiment may be beneficially used in other embodiments without specific recitation.

DESCRIPTION OF EXAMPLE EMBODIMENTS Overview

One embodiment presented in this disclosure is an electrical integrated circuit, comprising: a frequency comparator, connected to a ring oscillator and an external reference clock signal, configured to determine operational characteristics of an electrical circuit connected to an optical modulator based on a frequency difference between the ring oscillator and the external reference clock signal; an electrical driver connected with the optical modulator and configured to drive a phase shift of a first optical signal carried on a first arm of the optical modulator relative to a second optical signal carried on a second arm of the optical modulator, the electrical driver comprising: a first signal pathway, connected to the first arm of the optical modulator, wherein the first signal pathway includes: a first adjustable gain inverter, electrically connected to a first node and a second node on the first signal pathway; a first fixed gain inverter, electrically connected to the first node and the second node; a first inductor electrically connected between the second node and a third node on the first signal pathway; and a first non-inverting amplifier connected between the third node and the first node.

One embodiment presented in this disclosure is a method, comprising: monitoring a frequency of a ring oscillator of an electrical integrated circuit connected to an optical modulator to determine operational characteristics of the electrical integrated circuit; setting, based on the operational characteristics, a driving voltage used for an adjustable gain inverter, a fixed gain inverter, and a boost inverter that control the optical modulator; setting an amplification strength for the adjustable gain inverter based on the operational characteristics; and boosting an input signal to the fixed gain inverter and the adjustable gain inverter via the boost inverter.

One embodiment presented in this disclosure is a driver circuit, comprising: a plurality of buffers arranged in series, wherein a given buffer of the plurality of buffers is configured to accept a first input and a second input, and to produce a first output and a second output, and wherein each buffer of the plurality of buffers, comprises: a first adjustable gain inverter electrically connected to a first input node that is configured to receive the first input and to a first intermediate node; a first fixed gain inverter that is electrically connected in parallel with the first adjustable gain inverter to the first input node and to the first intermediate node; a second adjustable gain inverter electrically connected to a second input node that is configured to receive the second input and to a second intermediate node; a second fixed gain inverter connected in parallel with the second adjustable gain inverter; a first cross-coupled inverter that has an input connected to the first input node and an output connected to the second input node; a second cross-coupled inverter connected in parallel with the first cross-coupled inverter and having an output connected to the first input node and an input connected to the second input node; a first inductor connected between the first intermediate node and a first output node; a second inductor connected between the second intermediate node and a second output node; a first boost segment; and a second boost segment; wherein when the given buffer is not a last buffer in the series: the first boost segment includes a first non-inverting amplifier having an output connected to the first input node and an input connected to the first output node and the second boost segment includes a second non-inverting amplifier having an output connected to the second input node and an input connected to the second output node; and wherein when the given buffer is the last buffer in the series: the first boost segment includes a first inverting amplifier having an input and an output connected to the first input node and the second boost segment includes a second inverting amplifier having an input and an output connected to the second input node.

Example Embodiments

The present disclosure provides systems and methods for tuning an electrical driver of an optical modulator. The tuning parameters are based on the operational switching characteristics of the individual modulator connected with and driven by the electrical driver. Rather than setting the driving voltage and amplification characteristics of the electrical driver based on most-power-intensive operational characteristics for the optical modulator (e.g., the worst-case power consumption characteristics of a modulator still compliant with design parameters/specification for that type/class/part-number of modulator) regardless of actual operational characteristics of the optical modulator, the electrical driver described herein matches the driving voltage and amplification characteristics to how an individual optical modulator operates. By individualizing the driving voltage and amplification characteristics of an electrical driver, a fabricator of photonic platforms (or devices for use therein) may thus improve power consumption in less-power-intensive process corners (e.g., avoiding over supplying of voltage to individual components), increase signaling bandwidth availability, and boost process yields by expanding the operational characteristics that define the process corners that meet the design parameters/specifications for a class of modulators.

FIG. 1 is a block diagram of a photonic platform 100 that includes an electrical integrated circuit (EIC) 110 and an optical integrated circuit 120 (OIC). The EIC 110 and OIC 120 include different components that are fabricated using different processes and are joined together to provide electrical control and monitoring of optical signals as well as monitoring of the process corner behavior of the EIC 110.

The EIC 110 includes a frequency comparator 130, a ring oscillator 140, and an electrical driver 180 (generally, driver 180) that is connected to the OIC 120. In some embodiments, the EIC 110 includes a reference clock 150, a processor 160, and a programmable voltage regulator 170. In some embodiments, one or more of the reference clock, 150, the processor 160, and the regulator 170 may be provided externally to the EIC 110, such as by temporary connection to test equipment, or a connection with a second electrical circuit.

The OIC 120 includes an optical modulator 121, such as a Mach-Zehnder Interferometer (MZI). The OIC 120 may include one or several optical modulators 121, each associated with a connection to the driver 180, as well as other optical elements (and associated electrical interconnects) such as, but not limited to: waveguides, optical interfaces (e.g., to optically connect the OIC 120 to a second optical circuit or an optical cable), light detectors, light generators, lenses, reflectors, etc. The driver 180 is configured to drive a phase shift in one or more arms of the modulator 121 to shift the phase of a given optical signal carried by the modulator 121 relative to another optical signal carried by the modulator 121.

During self-test or external calibration, EIC 110 and the OIC 120 are connected, and the frequency comparator 130 compares the frequency of the ring oscillator 140 with an external reference clock signal from the reference clock 150 to determine the operational characteristics of the driver 180, including a frequency difference between two signals carried/supplied by the ring oscillator 140 and the reference clock 150. Depending on how the driver 180 is fabricated, the individual arms of the driver 180 may exhibit different operations characteristics, typically classified as Fast (denoted as F), Slow (denoted as S), or Typical (denoted as T) that indicate the duty cycle of an N or a P channel in the modulator 121. Fast corners exhibit carrier mobilities that are higher than Typical and Slow corners, and Slow corners exhibit carrier mobilities that are lower than Typical and Fast corners, which affects the speed at which a given transistor in the driver 180 is able to switch and the amount of power required to affect that transistor to switch.

The frequency comparator 130 provides a Process/Voltage/Temperature (PVT) readout to the processor 160, which may be burnt into the EIC 110 as a chip fuse value. The processor 160 interprets the frequency of the ring oscillator 140 to determine which process corner (e.g., FF, FS, TT, SF, or SS) the driver 180 belongs to, and how to set/tune the driver 180 to account for the process corner. The processor 160 sets the gain of one or more adjustable/tunable inverters in the buildup stage 182 of the driver 180 to account for the identified process corner of the driver 180, and sets the supply voltage of the regulator 170 accordingly. Setting the programmable drive of each tunable inverter in the buildup stage 182 controls for excessive peaking across the range of process variation in the driver 180, which protects the transistors from damage (e.g., due to over voltage) and reduces optical eye closure in the optical signals modulated by the modulator 121.

The regulator 170 provides a supply voltage (denoted herein as VDD) as a driving voltage and/or a reference voltage for various components of the driver 180 (including the input stage 181 and the buildup stage 182), which controls the strength and power consumption of the driver 180. By matching the power supplied to the process corner that an individual driver 180 exhibits, the regulator 170 may apply a more consistent amount of power to different process corners and use less power in Fast and Typical process corners than a regulator 170 configured to supply a power level based on the performance characteristics of a Slow process corner.

As illustrated, the driver 180 includes an input stage 181 and a buildup stage 182, although several input stages 181 and buildup stages 182 may be included to control several modulators 121 or elements thereof. The input stage 181 may include various pass through termination and differential amplification circuitry, such as the examples illustrated and discussed in greater detail in regard to FIGS. 5A and 5B. The buildup stage 182 amplifies and adjusts the strength of the differentiated signals produced in the input stage 181 and supplies the amplified signals 1901-n to the modulator 121. The number n of amplified signals 190 is based on the number of modulators 121 and individually driven stages thereof on the OIC 120. Example circuitry for a buildup stage 182 is discussed in greater detail in regard to FIGS. 6 and 7 .

FIG. 2A illustrates a multi-segment electro-optic modulation device, such as may be used as a modulator 121 according to embodiments described herein.

Specifically, FIG. 2A illustrates an MZI-type modulation device 200 that includes an input 205 that receives an optical signal (e.g., a continuous wave signal) and an output 245 that transmits an optical signal that is modulated based on the electrical signals that are used to drive optical modulator segments 220 _(1-n), 225 _(1-n). The input 205 includes a splitter 210 (e.g., a y-splitter) that splits an incoming continuous wave into each of branches 215 ₁, 215 ₂. The split optical signals each pass through various optical modulator segments 220 _(1-n), 225 _(1-n) as described above. The modulation segments 220, 225 of each branch 215 may be grouped into pairs 230, such that pair 230 ₁ corresponds to modulation segments 220 ₁ and 225 ₁, pair 230 ₂ to modulation segments 220 ₂ and 225 ₂, and so forth. In one embodiment, at least one of the pairs 230 is used for thermal biasing of the modulation device 200. For example, the modulation device 200 may be DC biased at the quadrature point by providing appropriate signals to the thermal biasing pair(s) to heat a portion of the modulation device 200 (referred to as a DC bias section (not illustrated). While n segments are illustrated for each branch 215 ₁, 215 ₂, differing numbers of modulating segments per branch are also possible. The modulated optical signals from each branch 215 are then recombined using a combiner 240 (e.g., a y-combiner) to form a modulated output optical signal at output 245. Moreover, the structure shown in modulation device 200 may be used in conjunction with any of the modulators and features disclosed herein.

FIG. 2B is a schematic view of a modulator segment pair 230 coupled with outputs of the modulator driver 180, according to embodiments described herein. Specifically, the view represents an electrical equivalent model of a pair 230 of modulator segments 220, 225 of a modulator 121 of a SISCAP (Silicon-Insulator-Silicon Capacitor) device.

Each modulator segment 220, 225 includes a resistance RP (respectively shown as resistors 270, 275) that represents the resistance of the P-type region of the SISCAP device, and a resistance RN (respectively shown as resistor 250, 255) representing the resistance of the N-type region. The resistances RP, RN of each modulator segment 220, 225 are coupled through a capacitance C (respectively shown as capacitors 260, 265), which generally represents the capacitance across a gate oxide, or gate dielectric of the SISCAP device (also referred to as a capacitive junction). In a SISCAP device, typically one of the N and the P plates of the capacitors 260, 265 is driven by the driver 180 at different voltages according to respective data signals 201A, 201B (generally, data signals 201), and the other plate is held at a constant DC voltage, V_(poly) (represented by connection 280). Although FIG. 2B illustrates that the P-type regions are held at V_(poly) and the N-type regions are driven at the variable voltage of the data signals 201, in other embodiments, the N-type regions may be held at V_(poly) and the P-type regions driven according to the data signals 201.

As shown, the electrical contacts of the N-type region are coupled at inputs 202A, 202B (generally, inputs 202) with a first arm 290A (generally, amplifier arm 290) and a second arm 290B of an amplifier chain of the driver 180 that that respectively provide a data signal 201A and a complement data signal 201B. Each amplifier arm 290 is driven at a shared voltage VDD and includes various inverters, inductors, Electro-Static Discharge (ESD) protection devices (also referred to as ESD protectors), which are discussed in greater detail in regard to FIGS. 6 and 7 herein. Each amplifier arm 290 receives the respective data signal 201A/201B and outputs an amplified data signal 203A, 203B (generally, amplified data signals 203) onto a corresponding input 202A, 202B of the pair 230. As such, the amplified data signals 203 generally represent the full rail-to-rail data signals 201, albeit at a different voltage that is matched to the capabilities of the SISCAP device. In embodiments using an even number of inverters, if data signal 201A corresponds to the data (DATA) and data signal 201B corresponds to the logical complement of the data (denoted as DĀTĀ, not(DATA), DATA*, etc.), then the amplified data signal 203A corresponds to the data and the amplified data signal 203B corresponds to the complement. In embodiments using an odd number of inverters, if data signal 201A corresponds to the data (DĀTĀ) and data signal 201B corresponds to the logical complement of the data (denoted as DĀTĀ, not(DATA), DATA*, etc.), then the amplified data signal 203A corresponds to the complement and the amplified data signal 203B corresponds to the data.

FIGS. 3A and 3B are block diagrams of a driver 180, according to embodiments described herein. Generally, driver arrangement 300 illustrates an example of the high-level functionality of a driver 180, and driver arrangement 350 illustrates one example implementation of the driver arrangement 300. Although the functionality of driver arrangement 300 is illustrated as a sequence of a plurality of functional stages, alternative embodiments may provide different sequences or different functional stages. For example, the functional stages (or portions of the stages) may be performed in a different order, may include additional stages or omit certain stages, or may perform certain functional stages contemporaneously. Additionally, various functional stages may be provided as separate integrated circuits, or some or all of the functional stages included within a shared integrated circuit.

As shown, driver arrangement 300 receives a differential input signal 302. Generally, differential signaling is a method of transmitting information electrically using two complementary signals sent on two paired wires or traces (called a differential pair). Because external signal interference, such as noise, tends to affect both wires in a similar manner, and because the transmitted information is contained in the difference between the complementary signals, differential signaling generally offers improved resilience to electromagnetic noise when compared with single-ended signaling. Generally, single-ended signaling transmits information using one signal relative to an unpaired reference voltage (e.g., ground). The noise immunity of differential signaling may be advantageous for use with low-voltage, high-speed electronics systems, such as modern communications systems. The differential input signal 302 may be an analog signal or digital.

In many cases, the modulator 121 may require particular signal characteristics at an input that do not match the characteristics of the data included in the differential input signal 302 (e.g., which may be dictated by a communications standard). In some embodiments, the modulation efficiency of the modulator 121 depends on a voltage output range provided by the circuitry of the driver 180 as well as where the operational characteristics of the modulator 121 fall within the production margins for that type of modulator 121. For example, a first compliant modulator 121 in a first process corner may require X mW (milliwatts) of power, whereas a second compliant modulator in a second process corner may require Y mW of power. By adjusting the supply voltage and inverter gains for particular process corners, rather than setting the driver 180 based on the requirements of one particular process corner, a fabricator may reduce the difference in the power consumption in opposite process corners (e.g., reducing Δ(X, Y)) due to process variations in fabricating the driver 180.

At the conversion stage 304, the differential input signal 302 is converted by driver circuitry to produce a complementary CMOS signal. Among other functions, conversion may include various stages of amplification of the differential input signal. In one embodiment, the conversion stage 304 provides a rail-to-rail complementary CMOS output signal. The information included in the complementary CMOS signal may be further processed in a processing stage 306, and the resulting signal may be amplified in an amplification stage 308 to generate a complementary CMOS output signal 310 suitable for driving a modulator. The amplification stage 308 may include a number of stages of amplifier circuitry that collectively boost the output signal into a desired voltage range for operating the modulator 121.

Driver arrangement 350 provides one example of the driver arrangement 300. A current-mode logic (CML) input signal 352 is received at a CML-to-CMOS conversion stage 354 of the driver arrangement 350. CML is one type of differential digital logic family, generally configured to transmit data at rates of up to 40 gigabits per second (Gb/s) and higher using printed circuit boards. CML is commonly used in the design of high-speed electronic systems due to a relatively small output voltage swing and a relative fast current switching. As with other logic families, CML may specify additional requirements, such as termination circuitry (discussed further below). In one embodiment, the CML-to-CMOS conversion stage 354 includes a differential amplifier whose output is coupled with at least one transimpedance amplifier (TIA). In one embodiment, the CML-to-CMOS conversion stage 354 includes one or more amplification stages following the TIA(s) to boost the signal to a rail-to-rail complementary CMOS output signal suitable for driving a CMOS-based modulator.

The output signal from the CML-to-CMOS conversion stage may have additional processing performed before being amplified to desired output levels for the modulator 121. As shown, a data multiplexing stage 356 allows the output signal data to be changed during operation. For example, the data multiplexing stage 356 may allow a test signal or other desired signal to be selected for output to the modulator 121. Following the data multiplexing stage 356, the signal is amplified at output amplification stage 358 to an output signal 360 having an output range suitable for driving the CMOS-based modulator.

FIG. 4 is a block diagram of an exemplary conversion stage of a modulator driver 180, according to embodiments described herein. The conversion stage 400 generally corresponds to the CML-to-CMOS conversion stage 354 in FIG. 3B.

The CML input signal 352 is received at differential inputs 402A, 402B (generally differential input 402). The differential signals 405P, 405N corresponding to the CML input signal 352 pass through termination circuitry 410 prior to being received at the differential amplification stage 415. For example, according to CML requirements, the termination circuitry 410 may include a 50 Ohm (0) resistor for each differential input 402, as well as a pull-up to a reference voltage (such as VDD). One example of termination circuitry 410 is provided in FIG. 5A.

The differential signals 405N, 405P are received at the differential amplification stage 415 and amplified. The output signals 420N, 420P from the differential amplification stage 415 are provided to a first inverter amplification stage 425. In one embodiment, the inverter amplification stage 425 includes one or more TIAs. In one embodiment, the inverter amplification stage 425 includes one TIA applied to each output signal 420N, 420P. Outputs of the first inverter amplification stage 425 are signals 430N and 430P. Each of the signals 430N, 430P is provided to a common-mode feedback stage 435 and to a chained and cross-coupled inverter amplification stage 445. Based on the signals 430N, 430P, the common-mode feedback stage 435 outputs a control signal 440 to the differential amplification stage 415. The control signal 440 is used to control the biasing of the inverter amplification stage 425 by modifying properties of the output signals 420N, 420P.

The chained and cross-coupled inverter amplification stage 445 includes one or more stages of coupling inverters, which include cross-coupled inverters. The output signals 450N, 450P from the chained and cross-coupled inverter amplification stage 445 are provided to outputs 452P, 452N and collectively are considered the output signal 450. In one embodiment, the output signal 450 from the conversion stage 400 is a rail-to-rail complementary CMOS signal, having an output range suitable for driving a CMOS-based modulator 121.

FIGS. 5A and 5B provide a schematic view of an input stage 181 of a driver 180, according to embodiments of the present disclosure. FIG. 5A illustrates a first portion of the input stage 181, generally corresponding to pass through termination circuitry 410. FIG. 5B generally corresponds to a differential amplification stage 415, an inverter amplification stage 425, and common-mode feedback stage 435 provided to a first inverter amplification stage 425.

FIG. 5A illustrates an embodiment of pass through termination circuitry as may be used in the termination circuitry 410. The pass through termination circuitry of the termination circuitry 410 receives the differential signals 405P, 405N and carries the differential output signals 405P, 405N on two arms to terminals A and B to the circuitry of the differential amplification stage 415, and example of which is illustrated in FIG. 5B.

The pass through termination circuitry 410 includes three sets of Electrostatic Discharge (ESD) protectors 540A-F (generally, ESD protectors 540) on each arm, which include a first diode 541 connected from ground to the arm, and a second diode 542 connected from the arm to a reference voltage, such as VDD.

Each arm includes an inductive pass-through voltage divider, such as a T-Coil structure, that includes a first inductor 550A, 550B in series/coupled with a second inductor 555A, 555B, and a pass-through bridge capacitor 560A, 560B parallel to the inductors. The first inductor 550A, 550B is connected to a first node 501A, 501B and a second node 502A, 502B respectively, and the second inductor 555A, 555B is connected to the second node 502A, 502B and a third node 503A, 503B respectively. The pass-through bridge capacitor 560A, 560B is connected at a first node 501A, 501B, and a third node 503A, 503B respectively.

Each arm runs to the first node 501A, 501B and from the second node 502A, 502B. A resistive voltage divider that includes a first dividing resistor 565A and a second dividing resistor 565B links the first arm with the second arm between the third nodes 503A, 503B. In various embodiments, the first dividing resistor 565A and the second dividing resistors 565B are of the same resistance, but different resistances may be used in other embodiments to define a different ratio between the two arms and one or more termination resistors (not illustrated) may be added to each arm to reduce back reflection of signals carried thereon.

A capacitance and inductance between the second ESD protector 540B and the third ESD protector 540C is represented in FIG. 5A as capacitor 570A and inductor 575A. Similarly, a capacitance and inductance between the fifth ESD protector 540E and the sixth ESD protector 540F is represented in FIG. 5A as capacitor 570B and inductor 575B.

A resistive voltage divider that includes a first dividing resistor 580A and a second dividing resistor 580B links terminal A and terminal B together after the third ESD protector 540C and the sixth ESD protector 540F. In various embodiments, the first dividing resistor 580A and the second dividing resistors 580B are of the same resistance, to define a reference voltage V_(CM) halfway between that observed at terminals A and B. In various embodiments, the common mode voltage V_(CM) is generated by a separate generator (not illustrated). The resistors create a high impedance connection to nodes A and B to set those nodes to the generated V_(CM).

Turning now to FIG. 5B, the differential signals 405P, 405N are received from the circuitry of the termination circuitry 410 from terminals A and B, respectively, at the circuitry providing the differential amplification stage 415. The differential amplification stage 415 includes a differential amplifier that includes a first pair of MOS devices (i.e., transistors 510A, 510B; generally, first pair of MOS devices 510) having a first conductivity type (e.g., n-channel), with each of the first pair of MOS devices 510 configured to receive one differential signal 405P or 405N with reference to a source 520. Generally, the first pair of MOS devices 510 may also be referred to as a differential pair. The differential pair generally offers some noise immunity and common mode rejection, which results in an increased input sensitivity that is suitable for low voltage input signals. The differential amplifier also includes a second pair of MOS devices (i.e., transistors 515A, 515B; generally, second pair of MOS devices 515) having a different conductivity type (e.g., p-channel). Each of the second pair of MOS devices generally operates as a current source load for a respective MOS device of the first pair. For example, transistor 515A is coupled with transistor 510A, and 515B with 510B. The differential amplifier outputs signals 420P, 420N, represent the common mode output of the differential pair that is provided at the respective drains of the transistors 510A, 510B.

The output signals 420P, 420N are received at a first inverter amplification stage 425. As shown, the first inverter amplification stage 425 includes a first TIA coupled with one output of the differential amplifier, and a second TIA coupled with the other output of the differential amplifier. Each TIA includes a respective inverter 530A, 530B (generally inverters 530) and feedback resistor 531A, 531B (generally feedback resistors 531). The first inverter amplification stage 425 is generally configured to produce a second amplified signal (i.e., signals 430P, 430N) that is based on the first amplified signal (i.e., signals 420N, 420P). Because the input and output of each TIA is coupled through the respective feedback resistor 531, the input and common mode output of the TIAs may be controlled to have voltage values close to a trip point of the respective inverters 530. Additionally, the circuitry for the first inverter amplification stage 425 may include a second inverter 532A, 532B (generally inverters 532) on each signal pathway. In embodiments that use an odd number of stages in the chained and cross-coupled inverter amplification stage 445 (discussed in greater detail in regard to FIGS. 6 and 7 ), the second inverters 532A, 532B may be omitted.

Although not illustrated in FIG. 5B, each of the inverters 530, 532, 536 may be powered using a shared driving voltage VDD.

In some embodiments, any voltage offsets in the differential output signals 420N, 420P are mitigated by measuring the common mode at the output of the TIAs, and by comparing the common mode with the trip point of an inverter amplifier. A feedback or control signal 440 based on the difference value may be provided to the current source loads of the differential amplifier (i.e., transistors 515A, 515B) to adjust the common mode output of the differential pair. Adjustments made by the control signal 440 to the differential pair common mode output are propagated to the input of the TIAs, which in turn adjusts the bias of the TIAs to follow the trip point of the inverters 536. In some embodiments, the common mode output of the TIAs is select to have voltage values close to the trip point of a successive inverter in the sequence (e.g., inverters 532A, 532B), which results in a large gain to be available at the successive inverter, with nearly full rail-to-rail CMOS output levels produced by the successive inverter.

As shown, the common-mode feedback stage 435 includes resistors 535A, 535B that are each coupled with the output of a respective TIA, as well as with an input of an inverter 536. The trace carrying the control signal 440 may include one or more components for filtering or conditioning the control signal 440. For example, the output of inverter 536 may have a capacitor connected with voltage source VDD in order to perform frequency compensation. In one embodiment, the (conditioned) control signal 440 is coupled with the gates of the p-channel transistors 515A, 515B. In one embodiment, the characteristics of resistors 535A, 535B and/or inverter 536 are selected to provide a desired control signal 440.

The signals 430N, 430P output by the arrangement illustrated in FIG. 5B may be provided to a buildup stage 182, such as that illustrated in FIG. 6 . FIG. 6 illustrates a schematic view of the buildup stage 182 including a chain of inverting buffers 630 _(1-n) and an inverting buffer 650 arranged in series across a first arm 610A and a second arm 610B with the inverting buffer 650 arranged as the last buffer in the series of buffers. An example schematic view of an individual inverting buffer 630 is provided in FIG. 7 . An example schematic view of the inverting buffer 650 is provided in FIG. 8 .

In FIG. 6 , the first arm 610A receives the signal 430P, which is provided to the respective first-arm stages of the inverting buffers 630 _(1-n), and the inverting buffer 650, which output signal 450P to the modulator 121. The second arm 610B in FIG. 6 receives the signal 430N, which is provided to the respective second-arm stages of the inverting buffers 630 _(1-n) and inverting buffer 650, which output signal 450N to the modulator 121.

In various embodiments, the first arm 610A and the second arm 610B include respective ESD protectors 640A, 640B between the inverting buffer 650 and the modulator 121. Each of the ESD protectors 640A, 640B includes a first diode 641A, 641B connected from ground to the respective arm, and a second diode 642A, 642B connected from the respective arm to a reference voltage VDD.

FIG. 7 illustrates a schematic view of an individual inverting buffer 630 in which the first arm 610A runs through a first node 715, a second node 725, and a third node 735 and the second arm 610B runs through a fourth node 745, a fifth node 755, and a sixth node 765. Successive inverting buffers 630 are connected such that the third node 735 of a prior inverting buffer 630 _(x) is connected to the first node 715 of a successive buffer 630 _(x+1) and the sixth node 765 of a prior inverting buffer 630 _(x) is connected to the fourth node 745 of a successive inverting buffer 630 _(x+1). Any number of inverting buffers 610 _(1-n) may thus be arranged serially in a chain arrangement. The first node 715 and the fourth node 745 of a first inverting buffer 610 ₁ are connected to the input stage 181 (e.g., to the respective second inverting inductors 532A, 532B). The third node 735 and the sixth node 765 of the last inverting buffer 630 n are connected to the inputs of the inverting buffer 650.

In the inverting buffer 630, a cross-coupled pair of inverters 710 and 720 are disposed between the first node 715 and the second node 725 to provide DC offset correction. The first cross-coupled inverter 710 and the second cross-coupled inverter 720 are oriented in opposite orientations relative to one another.

A first adjustable gain inverter 730A and a first fixed gain inverter 740A are disposed in parallel to one another between the first node 715 and the second node 725. Similarly, a second fixed gain inverter 740B and a second adjustable gain inverter 730B are disposed in parallel to one another between the fourth node 745 and the fifth node 755. In some embodiments, the gain of the adjustable gain inverters 730A, 730B of a next or subsequent inverting buffer 630 _(x+1) is higher than the gain of the adjustable gain inverters 730A, 730B in an earlier or prior inverting buffer 630 _(x) in the buildup stage 182 so that the strength of the signal is progressively made stronger across the buildup stage 182.

A first inductor 760A is disposed between the second node 725 and the third node 735, and a second inductor 760B is disposed between the fifth node 755 and the sixth node 765, which increases the available bandwidth for the signal carried through the inverting buffer 630 without consuming additional power.

A first non-inverting amplifier (NIA) 750A is disposed between the first node 715 and the third node 735, which may have and adjustable of fixed gain. Similarly, a second NIA 750B, with a similarly selected adjustable or fixed gain, is disposed between the fourth node 745 and the sixth node 765. Although illustrated as a single component, in various embodiments, an even number of inverters and/or more than one non-inverting amplifier can be included in the NIA 750. For example, a NIA 750 may include two inverters in series. Each NIA 750 is connected with an input side downstream of the adjustable gain inverter 730, fixed gain inverter 740, and inductor 760 to receive their output as an input, which is delayed, amplified and returned as an output that is combined with a later input to the adjustable gain inverter 730 and fixed gain inverter 740. The amount of delay introduced by the inductor 760 and the individual inverters in the NIA 750 can be calibrated based on the signaling and control needs of the buildup stage 182.

Each of the inverters 710, 720, 730, 740, and NIA 750 have the same driving voltage VDD applied by the programmable regulator 170. The value of the driving voltage VDD and the gain programmed for the adjustable gain inverters 730 and NIA 750 are based on the operational characteristics of the modulator 121 that the buildup stage 182 is driving. By adjusting the gain and VDD, a fabricator can mitigate the risk of excessive peaking within particular process corners, which can cause output optical eye closure in the modulator 121 and damage transistors in the electrical driver 180. The presently described driver 180 can thus drive signals for a wide variety of modulators 121 with a lower power consumption than comparable designs, because the driver 180 is optimized to the individual drive strength of a particular modulator 121 instead of the slow-slow process corner across all driver/modulator pairings as other drivers 180 are optimized for. This provides for a more consistent power consumptions across all process corners and does not consumer additional power in the typical or fast process corners.

FIG. 8 illustrates a schematic view of the inverting buffer 650 in which the first arm 610A runs through a first node 815, a second node 825, and a third node 835 and the second arm 610B runs through a fourth node 845, a fifth node 855, and a sixth node 865. The first node 815 and the fourth node 845 of the inverting buffer 650 are connected to the third node 735 and the sixth node 765 of the last inverting buffer 630 n. The third node 835 and the sixth node 865 of the inverting buffer 650 are connected to outputs of the driver 180. The inverting buffer 650 is positioned in the chain/series of buffers as the last buffer of the plurality of buffers.

In the inverting buffer 650, a cross-coupled pair of inverters 810 and 820 are disposed between the first node 815 and the second node 825 to provide DC offset correction. The first cross-coupled inverter 810 and the second cross-coupled inverter 820 are oriented in opposite orientations relative to one another.

A first adjustable gain inverter 830A and a first fixed gain inverter 840A are disposed in parallel to one another between the first node 815 and the second node 825. Similarly, a second fixed gain inverter 840B and a second adjustable gain inverter 830B are disposed in parallel to one another between the fourth node 845 and the fifth node 855. In some embodiments, the gain of the adjustable gain inverters 830A, 830B is higher than the gain of the adjustable gain inverters 730A, 730B in the last inverting buffer 630 _(n) in the buildup stage 182 so that the strength of the signal is progressively made stronger across the buildup stage 182.

A first inductor 860A is disposed between the second node 825 and the third node 835, and a second inductor 860B is disposed between the fifth node 855 and the sixth node 865, which increases the available bandwidth for the signal carried through the inverting buffer 630 without consuming additional power.

In contrast to the non-inverting buffers 630, the inverting buffer 650 does not include NIAs 750 between the first node 815 and the third node 835 or the fourth node 845 and the sixth node 865. Rather, the inverting buffer 650 includes a first inverting amplifier 850A on the first arm 610A at the first node 815 and a second inverting amplifier 850B on the second arm 610B at the fourth node 845. The input and the output sides of the first inverting amplifier 850A are both connected to the first node 815, and the input and the output sides of the second inverting amplifier 850B are both connected to the fourth node 845. Although illustrated as a single component, in various embodiments, any odd number of inverters can be included in the inverting amplifier 850. For example, an inverting amplifier 850 may include three inverters in series. In some embodiments, the inverters in the inverting amplifier 850 can be fixed gain or adjustable gain.

Each of the inverters 810, 820, 830, 840, and inverting amplifiers 850 have the same driving voltage VDD applied by the programmable regulator 170. The value of the driving voltage VDD and the gain programmed for the adjustable gain inverters 830 and inverting amplifiers 850 are based on the operational characteristics of the modulator 121 that the buildup stage 182 is driving. By adjusting the gain and VDD, a fabricator can mitigate the risk of excessive peaking within particular process corners, which can cause output optical eye closure in the modulator 121 and damage transistors in the electrical driver 180. The presently described driver 180 can thus drive signals for a wide variety of modulators 121 with a lower power consumption than comparable designs, because the driver 180 is optimized to the individual drive strength of a particular modulator 121 instead of the slow-slow process corner across all driver/modulator pairings as other drivers 180 are optimized for. This provides for a more consistent power consumption across all process corners and does not consume additional power in the typical or fast process corners.

As used herein, the NIA 750 of the inverting buffers 630 and the inverting amplifiers 850 of the inverting buffers 650 may collectively be referred to as boost segments for the buffers, and the inverters therein may be referred to as the boost inverters.

FIG. 9A illustrates a first eye diagram 900 a (generally, eye diagram 900) and FIG. 9B illustrates a second eye diagram 900 b, according to embodiments of the present disclosure. An eye diagram 900 is a histogram that illustrates the amplitude of one or more signals 920 a-b (which may include the complements to those signals) that superimposes the amplitudes of those signals over a signaling interval (shown here from times t₁ to t₁₆) to illustrate various characteristics of the signals based on the appearance and pattern of an “eye pattern” in the eye diagram 900. The illustrated eye diagrams 900 a-b include two levels of modulation for values between a high rail 910 a and a low rail 910 b (generally, rail 910). Other modulation schemes, such as for a Pulse Amplitude Modulation signal with four, eight, sixteen, etc., levels of pulse modulation (also referred to as PAM4, PAM8, PAM16, etc.) may include more than two rails 910.

Gaps in the signals 920 a-b define an eye 930 a whose size and position relative to the rails 910 over the time period indicate the effects of noise, signal degradation, signal path interruption, synchronization, jitter, and the like. Differences between the first eye diagram 900 a and the second eye diagram 900 b illustrate an increase in the bandwidth of the signals 920 as may be provided by the present disclosure by boosting the transition-rate of the signals 920 to reach equilibrium values 940 a-b (generally, equilibrium value 940) faster, which is smaller in amplitude than the rail 910 (as in FIG. 9B) than if the signals 920 were un-boosted (as in FIG. 9A). The non-inverting amplifiers 750 and the inverting amplifiers 850 provide the boost to the signals 920 carried through the buildup stage 182 that allows the signals 920 to reach the equilibrium value 940 faster. Accordingly, the boost segment enables the eye 930 to “open” faster, which results in higher signal bandwidth.

FIG. 10 is a flowchart of a method 1000 for tuning a driver 180 for a modulator 121. Method 1000 beings at block 1010, where a processor 160 monitors a frequency of a ring oscillator 140 connected to the driver 180 to determine operational characteristics of the driver 180. In various embodiments, the operational characteristics include identifying a process corner that the driver 180 belongs to (e.g., Fast-Fast, Slow-Slow, Typical-Typical).

At block 1020, the processor 160 sets a voltage regulator 170 to provide a reference voltage VDD to the driver 180 at an appropriate voltage level based on the determined operational characteristics of the driver 180. In various embodiments, the voltage VDD is used as a driving voltage or a reference voltage for various components in the driver 180 including ESD protectors, inverters (amplifying and cross-coupling), and transistors.

At block 1030, the processor 160 sets an amplification strength of a plurality of tunable inverters (i.e., adjustable gain) included in a buildup stage 182 of the driver 180. In various embodiments, each signal of a differentiated signal is carried on a separate arm of the driver 180, and the tunable inverters on each arm are tuned separately based on the operation characteristics of the driver 180 to which the given arm of the driver 180 is connected. The tunable inverters are arranged in parallel with a paired non-tunable inverter (i.e., fixed gain), and several pairs of inverters are arranged in series (with progressively larger strengths) within the buildup stage 182.

At block 1040, the processor 160 calibrates signal boosting in the boost segments of the buildup stage 182. Signal boosting through the boost segments (e.g., the NIA 750 and inverting amplifiers 850) imparts a delay to the signal carried through the boost segment in question as well as signal amplification, which produces a delayed output signal that is combined with a subsequent input to the fixed and adjustable gain inverters for that buffer. A fabricator may adjust the amount of delay in the boost segments (e.g., the timing of the signal input to the boost segments versus the timing of the signal output from the boost segments) based on the number of inverters included in the boost segment, a presence/strength of an analog delay line, etc.

Method 1000 may then conclude.

In the current disclosure, reference is made to various embodiments. However, the scope of the present disclosure is not limited to specific described embodiments. Instead, any combination of the described features and elements, whether related to different embodiments or not, is contemplated to implement and practice contemplated embodiments. Additionally, when elements of the embodiments are described in the form of “at least one of A and B,” it will be understood that embodiments including element A exclusively, including element B exclusively, and including element A and B are each contemplated. Furthermore, although some embodiments disclosed herein may achieve advantages over other possible solutions or over the prior art, whether or not a particular advantage is achieved by a given embodiment is not limiting of the scope of the present disclosure. Thus, the aspects, features, embodiments and advantages disclosed herein are merely illustrative and are not considered elements or limitations of the appended claims except where explicitly recited in a claim(s). Likewise, reference to “the invention” shall not be construed as a generalization of any inventive subject matter disclosed herein and shall not be considered to be an element or limitation of the appended claims except where explicitly recited in a claim(s).

The flowchart illustrations and block diagrams in the Figures illustrate the architecture, functionality, and operation of possible implementations of systems, methods, and computer program products according to various embodiments. In this regard, each block in the flowchart illustrations or block diagrams may represent a module, segment, or portion of code, which comprises one or more executable instructions for implementing the specified logical function(s). It should also be noted that, in some alternative implementations, the functions noted in the block may occur out of the order noted in the Figures. For example, two blocks shown in succession may, in fact, be executed substantially concurrently, or the blocks may sometimes be executed in the reverse order, depending upon the functionality involved. It will also be noted that each block of the block diagrams and/or flowchart illustrations, and combinations of blocks in the block diagrams and/or flowchart illustrations, can be implemented by special purpose hardware-based systems that perform the specified functions or acts, or combinations of special purpose hardware and computer instructions.

In view of the foregoing, the scope of the present disclosure is determined by the claims that follow. 

We claim:
 1. An electrical integrated circuit, comprising: a frequency comparator configured to determine operational characteristics of an electrical circuit; and an electrical driver configured to drive a phase shift of a first optical signal carried on a first arm of an optical modulator relative to a second optical signal carried on a second arm of the optical modulator.
 2. The electrical integrated circuit of claim 1, wherein the electrical driver comprises: a first signal pathway, connected to the first arm of the optical modulator, wherein the first signal pathway includes: a first adjustable gain inverter, electrically connected to a first node and a second node on the first signal pathway; a first fixed gain inverter, electrically connected to the first node and the second node; a first inductor electrically connected between the second node and a third node on the first signal pathway; and a first non-inverting amplifier connected between the third node and the first node.
 3. The electrical integrated circuit of claim 2, further comprising: a second signal pathway, connected to a second junction of the second arm of the optical modulator, wherein the second signal pathway includes: a second adjustable gain inverter, electrically connected to a fourth node and a fifth node on the second signal pathway; and a second fixed gain inverter, electrically connected to the fourth node and the fifth node; a second inductor electrically connected between the fifth node and a sixth node on the second signal pathway; and a second non-inverting amplifier connected between the sixth node and the fourth node.
 4. The electrical integrated circuit of claim 3, wherein a first gain of the first adjustable gain inverter and a second gain of the second adjustable gain inverter are set based on the operational characteristics.
 5. The electrical integrated circuit of claim 3, wherein the first signal pathway carries a first signal to the first arm of the optical modulator and the second signal pathway carries a logical complement of the first signal to the second arm of the optical modulator.
 6. The electrical integrated circuit of claim 5, wherein the first signal is applied to a first capacitive junction of the first arm of the optical modulator and the logical complement is applied to a first capacitive junction of the second arm of the optical modulator, wherein a second capacitive junction of the first arm and a second capacitive junction of the second arm are held at a constant voltage.
 7. The electrical integrated circuit of claim 3, further comprising: a cross-coupling, including: a first coupling inverter having a first input electrically connected to the first node and a first output electrically connected to the fourth node; and a second coupling inverter having a second output electrically connected to the first node and a second input electrically connected to the fourth node.
 8. The electrical integrated circuit of claim 7, further comprising a voltage regulator configured to produce a driving voltage that is set based on the operational characteristics and configured to apply the driving voltage to the first adjustable gain inverter, the second adjustable gain inverter, the first fixed gain inverter, the second fixed gain inverter, the first coupling inverter, the second coupling inverter, the first non-inverting amplifier, and the second non-inverting amplifier.
 9. The electrical integrated circuit of claim 2, wherein the first signal pathway includes: a second adjustable gain inverter electrically connected to a fourth node and a fifth node on the first signal pathway; a second fixed gain inverter electrically connected to the fourth node and the fifth node; a second inductor electrically connected between the fifth node and a sixth node on the first signal pathway; and an inverting amplifier connected at an input side and at an output side to the fourth node.
 10. The electrical integrated circuit of claim 9, wherein the first non-inverting amplifier includes an even number of inverters arranged in series and configured to receive input from the third node and provide output to the first node.
 11. The electrical integrated circuit of claim 2, wherein the first signal pathway further comprises: an electrostatic discharge protector attached between a third node and the optical modulator.
 12. A method, comprising: monitoring operational characteristics of an electrical integrated circuit; setting, based on the operational characteristics, a driving voltage used for an adjustable gain inverter, a fixed gain inverter, and a boost inverter that control an optical modulator; setting an amplification strength for the adjustable gain inverter based on the operational characteristics; and boosting an input signal to the fixed gain inverter and the adjustable gain inverter via the boost inverter.
 13. The method of claim 12, wherein an input of the adjustable gain inverter is connected with an input of the fixed gain inverter; wherein an output of the adjustable gain inverter is connected with an output of the fixed gain inverter; wherein an output of the boost inverter is connected to the input of the fixed gain inverter; and wherein boosting the input further comprises: delaying the input signal to a corresponding boost inverter; amplifying the input signal to produce a delayed output signal; and combining the delayed output signal with a next input signal at the input of the fixed gain inverter and the adjustable gain inverter.
 14. The method of claim 12, wherein setting an amplification strength for the adjustable gain inverter applies a stronger amplification to a subsequent adjustable gain inverter in a plurality of adjustable gain inverters relative to prior adjustable gain inverters of the plurality of adjustable gain inverters when the plurality of adjustable gain inverters are arranged in a chain.
 15. The method of claim 12, wherein the operational characteristics indicate a process corner of an electrical driver to which the adjustable gain inverter and the fixed gain inverter belong.
 16. A driver circuit, comprising: a plurality of buffers arranged in series, wherein a given buffer of the plurality of buffers is configured to accept a first input and a second input, and to produce a first output and a second output, wherein when the given buffer is not a last buffer in the series: a first boost segment includes a first non-inverting amplifier having an output connected to a first input node and an input connected to a first output node and a second boost segment includes a second non-inverting amplifier having an output connected to a second input node and an input connected to a second output node; and wherein when the given buffer is the last buffer in the series: the first boost segment includes a first inverting amplifier having an input and an output connected to the first input node and the second boost segment includes a second inverting amplifier having an input and an output connected to the second input node.
 17. The driver circuit of claim 16, wherein each buffer of the plurality of buffers, comprises: a first adjustable gain inverter electrically connected to a first input node that is configured to receive the first input and to a first intermediate node; a first fixed gain inverter that is electrically connected in parallel with the first adjustable gain inverter to the first input node and to the first intermediate node; a second adjustable gain inverter electrically connected to a second input node that is configured to receive the second input and to a second intermediate node; a second fixed gain inverter connected in parallel with the second adjustable gain inverter; a first cross-coupled inverter that has an input connected to the first input node and an output connected to the second input node; a second cross-coupled inverter connected in parallel with the first cross-coupled inverter and having an output connected to the first input node and an input connected to the second input node; a first inductor connected between the first intermediate node and a first output node; a second inductor connected between the second intermediate node and a second output node; a first boost segment; and a second boost segment.
 18. The driver circuit of claim 17, wherein the first adjustable gain inverter and the second adjustable gain inverter of subsequent buffers in the plurality of buffers are tuned to greater gains than the first adjustable gain inverter and the second adjustable gain inverter of earlier buffers in the plurality of buffers.
 19. The driver circuit of claim 17, wherein the last buffer of the plurality of buffers is connected to an optical modulator via a first arm and a second arm; wherein the first arm comprises a first electrostatic discharge protector; and wherein the second arm comprises a second electrostatic discharge protector.
 20. The driver circuit of claim 17, wherein the first non-inverting amplifier and the second non-inverting amplifier each include an even number of inverters arranged in series, and wherein the first inverting amplifier and the second inverting amplifier each include an odd number of inverters arrange in series. 